My own purpose in developing a 10-megabit radio link was threefold: to explore the design techniques underlying high-bandwidth microwave communications; to do my part to help bring Amateur Radio into the twenty-first century; and, most important of all, to use my company's T-1 Internet service at home for free. :-) This page is intended to describe my work to fellow Amateurs and other experimenters for whom a 9600-bps packet BBS or 31-baud PSK rig just doesn't cut it.
I've tried to describe the circuit and relevant construction techniques in depth, for those who may be interested in reproducing and/or modifying my work, but it should be made clear at the outset that such a project isn't a weekend endeavor for the electronics neophyte. While this isn't the most complex or demanding of projects, it has several unforgiving aspects that will not tolerate sloppy construction practices or inadequate care on the builder's part. You should be conversant with basic RF theory and construction skills before tackling it, or at least have access to one or two Elmers who have "been there and done that." You must have access to some basic electronic test equipment, including at a minimum a good oscilloscope and digital VOM. Finally, you should be prepared to commit the necessary resources of time and money to get the results you want. Underestimating the effort and expense required to get your project up and running is probably the #1 cause of failure.
Although dish antennas are clearly the best choice when maximum range
is the goal, they aren't likely to be necessary for links which span less
than a mile (1.6 km) or so. The prototype transceivers use homemade horn
antennas fabricated according to a 1:1-scale template generated by Paul
Wade, N1BWT's
excellent HDL_ANT antenna-design program. Here are two ready-to-build designs
suitable for use with the M/A-Com 10 GHz
and 24 GHz Gunn transceivers, respectively.
Initial testing and alignment of the link can be accomplished with no antennas
at all, but unless you are contemplating extremely close-range work (less
than a hundred feet or so with no obstructions) you will want to fabricate
horn antennas for your transceivers. You should observe about 5 dBi (i.e.,
5 dB of gain over an isotropic radiator in free space) with the "naked"
Gunnplexer apertures alone, and about 8-10 dB of additional gain with the
15 dBi 10 GHz horn mentioned above. To construct each horn as shown in
the photo, I used sheet brass obtained
from a local hobby shop, scoring the inner seams with a Dremel tool to
allow the antenna to be folded together origami-style, and soldered the
remaining seam. There were plenty of leftover scraps available for mounting
the antenna to the Gunn transceiver aperture. It's important to achieve
a tight fit between the antenna and aperture -- don't let the small end
of the horn "float" too far off the Gunnplexer's mounting surface. N1BWT's
latest page has a great deal of
knowledge in store for those who are interested in optimizing microwave
antenna performance, especially his online Microwave
Antenna Book.
The financially well-endowed may choose to go with a commercial standard-gain
horn antenna such as the HWR-90
model sold by Microtech, Inc.,
but their price of U.S. $400 each will likely send most experimenters reaching
for their dust mask, safety goggles, and Dremel tool instead.
In addition to the 10 GHz devices mentioned above, SHF Microwave also
distributes 24 GHz Gunn transceivers. These units are essentially identical
to the 10 GHz devices used in the link, and can be substituted with no
circuit modifications. They are less expensive as well. However, they are
quite a bit smaller and somewhat less powerful (5 mW versus 10 mW) than
their 10 GHz counterparts. I used a pair of the 24 GHz Gunnplexers in my
initial experiments while developing the radio link earlier this year,
but found that the signal they emitted was attenuated too severely by the
glass window at the "office" end of the link. The 10 GHz Gunnplexers had
no problem putting a good signal through the glass, especially after I
upgraded them to the 100 mW Gunn diodes! Not being especially talented
at metalworking, I also found it easier to fabricate the larger horn antennas
for the 10 GHz units. Finally, there are plenty of reasonably-inexpensive
spectrum analyzers available on the surplus market, including my own Tektronix
492, that make it trivial to set the 10 GHz Gunnplexers to any desired
frequency. 24 GHz spectrum analyzers,
on the other hand, still seem to cost about as much as a new Porsche. All
in all, I prefer the 10 GHz Gunnplexers to the 24 GHz parts and recommend
them as a first choice if you don't have any particular reason for wanting
to run your link on 24 GHz. From what Alan is reporting, it appears that
both models will continue to be available in the foreseeable future.
Data sheets for the 10 GHz and 24
GHz Gunnplexers are available here. Alan has annotated the 24 GHz model's
data sheet to clarify the misleading pin nomenclature used by M/A-Com.
As with any sensitive microwave semiconductor device, it's vital to keep
static electricity away from the various Gunnplexer diodes, especially
the mixer diode. These units come with shorting straps that should not
be removed until you've soldered the 2.2K mixer load resistor into place
between the mixer diode and ground terminals. Don't apply power until you've
checked and double-checked your pin connections!
I will attempt to keep this page up to date with any new information
that comes to light on the availability of suitable transceiver modules.
Please email me if you're aware of any
other sources for Gunn transceivers. (The cheaper surplus models that don't
include varactor diodes are not suitable for use at 10 megabits per second,
as they lack the necessary high-speed FM modulation capabilities.)
While the Mini-Circuits filters are relatively inexpensive, they could
easily be replaced with homemade filters similar to N6GN's, if you have
the equipment to test them. Several basic filter-design programs can be
found on the net, including Bob Lombardi's FilDes
for MS-DOS.
One important distinction between the IF strip shown here and the N6GN
design is the presence of a Mini-Circuits T4-1
broadband transformer between the Gunnplexer's mixer diode and the first
MAR-6 amplifier. The purpose of this transformer is to provide a better
match between the relatively high-impedance mixer diode and the 50-ohm
MMIC input. Without it, the effective receiver noise figure is degraded
by several dB due to excessive mixer-diode loading. The difference in weak-signal
recovery performance is dramatic. It's surprising that the original N6GN
article didn't include this useful tweak.
Besides the MC13155, the other key component in the FSK demodulator
is the quadrature tank circuit. This is a parallel LC resonant circuit
(visible to the right of the MC13155 IC in the picture)
which is tuned to the 145 MHz IF center frequency. As the FSK-modulated
IF signal varies above and below the center frequency, the quadrature tank
circuit exhibits a varying amount of phase shift which the MC13155 interprets
to determine the correct output logic level. Because the 10-Mbit/s data
rate requires a high signal bandwidth
for efficient transmission, it's necessary to deliberately lower the Q
factor of the quadrature tank circuit to approximately 3.6 by "swamping"
it with a 180-ohm resistor. The resulting demodulation bandwidth is given
by (145 / 3.6), or approximately 40 MHz. A signal near 125 MHz will yield
a low output level from the MC13155, while a signal near 165 MHz will result
in a high output level. The center IF frequency of 145 MHz corresponds
to the baseline Ethernet signal level which is present when no data is
being transmitted.
Guidelines for alignment and testing of the FSK demodulator circuit
appear below.
The filtered signal is then applied to a second OPA603 for amplification
to a level which is compatible with the Ethernet
AUI port. Note that this is still an "analog" signal in every sense of
the word -- it has not passed through any comparators or logic gates on
either the transmitter or the receiver side of the link. Keeping the Ethernet
signal in the analog realm by processing it with linear circuitry ensures
that we are able to preserve all of the edge characteristics of the signal
as it transitions to and from the "neutral" state at the beginning and
end of each packet. Experience has shown that Ethernet devices can be finicky
about edge-preservation, so our intention is to avoid potential edge-handling
complications by leaving the job of digital signal conditioning up to the
Ethernet hardware itself.
The output buffer circuit is fairly straightforward, with the exception
of the 938-ohm resistor that controls the DC offset at the output of the
first OPA603 op-amp. As the schematic's notes suggest, this resistance
value can be obtained with a parallel combination of 1K and 15K resistors,
or with a 10-turn precision trimmer. In either case, a good digital VOM
can help you obtain a resistance value close to 938 ohms, either by hand-selecting
1K and 15K resistors which yield a total resistance within 1% of the desired
value or by presetting the trimmer adjustment. The idea is to position
the output Ethernet signal at pin 6 of the final OPA603 to a baseline value
close to 0V DC when no signal modulation is present. Keeping the average
DC voltage near 0V helps avoid core saturation in the Mini-Circuits T1-1
broadband transformer which is used for isolation between the transceiver
and Ethernet device. Small changes in the 938-ohm resistor's value can
yield large shifts in the average DC output level, so it makes sense to
strive for precision in this part of the circuit. You may wish to verify
the DC offset of the output waveform with an oscilloscope, to make sure
the baseline level is within 100 mV or so of 0 VDC. The same unmodulated
145-MHz signal source used to align the FSK demodulator (below) can make
the output buffer offset easy to tweak as well.
In any event, you should resist the temptation to use AC coupling in
lieu of the DC level-shifting components in either the output buffer or
FSK modulator circuits. Ethernet packet traffic appears as a series of
periodic signal bursts with widely-varying duty cycles. Capacitors used
for AC coupling between stages with different DC operating points (such
as between the MC13155 and the output buffer amplifier, or between the
AUI transmit circuit and the Gunnplexer varactor diode) inevitably introduce
an RC time constant characteristic which can cause the average DC signal
level to vary over time, both between and during packet bursts. Because
of the aforementioned sensitivity of Ethernet hardware to the integrity
of signal edges at the beginning and end of each packet, any component
which introduces a time-variant DC characteristic is bound to be a bad
thing.
In fact, this effect cost me several weeks of head-scratching and an
astonishing amount of money during development of the prototype link. I
found I could receive either low-duty-cycle pings or continuous
high-intensity traffic near 10 Mbits/s, depending on the AC-coupled circuit
characteristics -- but I was never able to find a configuration that would
allow both types of traffic to pass! Given the complexities of choosing
the filters, amplifiers, signal levels, deviation and bandwidth figures,
and other operating conditions at both ends of the link, this was one problem
I did not need. (Most of the R&D budget for the project went straight
to Tektronix for one of their nifty new digital-phosphor
oscilloscopes, which, in desperation, I convinced myself I needed in
order to track down the link's signal-integrity problems. :-)
It should also be noted here that the AUI ports on the Ethernet hubs
I used were not enabled by default. Some experimentation revealed that
the hubs would not recognize an attached AUI transceiver unless all of
the ground pins on the AUI port were tied together to signal the presence
of a device plugged into the port. As the schematic indicates, pins 1,
8, 11, and 14 should be bonded together at the AUI plug. Because the AUI
signals themselves are differential in nature, this ground connection does
not need to be connected to the transceiver ground at either end. I have
heard vague rumors that some manufacturers' AUI ports also expect to see
at least a few milliamps' worth of loading at their +12V output pins in
order to recognize the presence of a connected transceiver device. This
was not the case with the LinkSys
hubs I used, but may be worth investigating further if you are unable to
get the link working in your case. It is easy enough to verify AUI-to-AUI
connectivity with a 4-wire crossover cable (pins 3-5 and 10-12 joined at
both ends) if you have doubts about your ability to communicate with your
particular hubs or NICs. Please email me
if you are able to verify or dispute any of these observations with respect
to the AUI interface.
The Ethernet signal to be transmitted enters the modulator circuit through
a Mini-Circuits T2-1T
broadband transformer, configured as a balun to provide isolation and an
improved impedance match between the 78-ohm AUI port and the Mini-Circuits
PLP-21.4
low-pass filter which follows it. By Carson's
Rule, the bandwidth occupied by a transmitted FM signal is related
to the highest modulating frequency present. Consequently, the low-pass
filter is employed to eliminate high-frequency energy
present in the Ethernet waveform which would otherwise waste bandwidth
without carrying useful information.
After filtering, the Ethernet signal is amplified by a Burr-Brown OPA603
wideband current-feedback op-amp. The variable resistor R1 controls the
amplifier's gain, which determines the amplitude of the output waveform
applied to the Gunn varactor diode and, consequently, the deviation
of the transmitted FSK signal, or the magnitude of its excursions from
its center frequency. At the OPA603's output, a 1N4148 diode provides cheap
insurance against possible damage to the Gunnplexer varactor caused by
forward-bias transients when powering the link up or down.
When designing a frequency-shift-keyed transmitter such as this one,
the desired FM deviation is typically chosen on the basis of the highest
modulating frequency expected to be transmitted. The deviation ratio,
given by the ratio of these two values, is not particularly critical, as
long as it is sufficient to generate intelligence-bearing sidebands of
reasonable amplitude, without occupying an unnecessarily-large chunk of
bandwidth. A deviation ratio between 1 and 2 is typical for such applications.
One reason the exact deviation ratio isn't critical is that a precise,
optimal transmission bandwidth budget is tricky to calculate for a complex
waveform such as an Ethernet signal, or even a voice transmission. Prior
to being used to modulate an FM transmitter, most of the energy in a 10-Mbit/s
Manchester-encoded baseband Ethernet signal exists around the 10 MHz point,
falling off several megahertz to either side. Very little signal-bearing
energy remains by the time our 21.4-MHz low-pass filter cutoff frequency
is reached, so the filter's purpose is largely to eliminate harmonics generated
in the process of forming the Ethernet waveform.
For those interested in a more detailed look at the subject, W6QS has
posted an excellent, easy-to-follow graphical model
of 10-Mbit/s Ethernet and its FSK modulation characteristics at the Tech
Bench Elmers Amateur Radio Society page.
Finally, an actual spectrum-analyzer screen shot of the FM-modulated
10 GHz signal from one of the prototype transceivers appears here.
This particular snapshot was taken while several megabytes' worth of MP3
files were being copied from one end of the link to the other at full Ethernet
speed.
Fortunately, because the relative frequency separation between the two
transceiver units is more important than their absolute frequencies, any
such drift-compensation circuit needs to be implemented at only one end
of the link, not both. In our case, the output buffer circuit at the "low-frequency"
(10.178 GHz) end of the link has a few extra parts. A low-gain amplifier
consisting of one stage from an LM324
quad op-amp is connected to monitor the differential outputs from pins
4 and 5 of the MC13155 FSK demodulator, with its output fed via a low-pass
RC integrating filter and bipolar transistor buffer to a voltage-summing
junction in the FSK modulator circuit. The OPA603 in the FSK modulator
circuit adds the constant voltage at its inverting input to the output
signal it delivers to the Gunnplexer varactor. (Because the sign of the
added voltage is inverted by the OPA603, we apply a negative voltage to
the summing junction to yield a positive bias offset at the varactor.)
The result is a simple feedback loop, in which any difference in the
DC level between the two FSK demodulator output pins is amplified, integrated
over time, and used to steer the IF difference frequency in the proper
direction to cancel out the unwanted offset voltage. The filter circuit,
consisting of a 4.7 uF capacitor and a 10K series resistor, is necessary
to keep the AFC loop from trying to cancel out the desired rapid changes
in frequency arising from the normal Ethernet modulation process.
At the other end of the link -- the "high-frequency" end which operates
at 10.323 GHz -- we simply feed the modulator summing junction with -8
volts through a fixed 820-ohm resistor, resulting in a constant DC offset
of about +5V. This is enough to bias the varactor diode into a reasonably-linear
portion of its voltage-versus-frequency characteristic curve, while providing
sufficient headroom on both sides for the applied Ethernet modulation.
If the Gunn varactor diode or its associated driving circuitry exhibits
any drift, the AFC circuit at the other end of the link will adjust its
own Gunnplexer's center frequency to compensate. The distinction between
the two transceivers is arbitrary -- note that we could just as easily
have applied AFC correction at the high-frequency transceiver unit instead,
swapping the connections between the MC13155 and LM324 to change the sign
of the correction signal.
With one transceiver mounted indoors and kept at a relatively constant
temperature of 75 degrees F, this AFC mechanism has proven capable of maintaining
the integrity of the link at Gunnplexer temperatures between approximately
50 and 140 degrees F as measured at the outdoor end of the link. Beyond
140 F, the system ran out of compensation range and failed. The excessive
heat buildup was due to the effect of the outdoor unit's near-airtight
RF shielding on a particularly warm day, aggravated by the use of plastic
wrap for rainproofing. A small fan was installed in the transceiver's outer
housing to aid ventilation, eliminating further problems.
In the prototype unit, this meter actually serves as a power-supply
current monitor in addition to monitoring the AFC circuit. The meter, installed
at the link control box, uses the 15' power-supply
ground line as its DC signal path return. Instead of a conventional zero-center
tuning meter as a frequency indicator, I used a 0-100 uA full-scale meter,
with a separate sensitivity control that adjusts the meter for a center-scale
reading under normal power-supply load conditions. Any fault condition
with the Gunn diode or other portion of the system with substantial current
requirements will deflect the meter off-center. This dual-purpose monitoring
scheme was a good match for the full-scale 0-100 uA meter I had on hand,
since the current drain from the optional 100 mW Gunn diode I used was
sufficient to drive the meter to center scale under normal operating conditons.
In reproducing the project, you may elect to use a standard zero-center
tuning meter with a separate signal ground line, or even omit the meter
circuit entirely.
The prototype transceivers use a pair of Extech
390GVS2 meters, available from Future-Active
Electronics. This digital panel meter is designed for direct readout
of applied voltages between 0 and 200 mV, so a resistive voltage divider
is used to bring the amplified voltage from the S-meter buffer back down
to the meter's display range. The resistor values shown will yield a displayed
voltage within a few millivolts of the actual RSSI output from the FSK
demodulator subassembly.
In one prototype unit, the following S-meter values were observed and
charted next to the actual voltage measured at the FSK demodulator subassembly's
RSSI output connection. The negative S-meter readings at low RF levels
are due to a combination of the LM324 buffer's offset voltage and the voltage
drop along the transceiver power cable's ground lead, the latter being
dominated by the substantial negative current consumed by the Gunn diode.
The input signal was an unmodulated 145.0 MHz carrier applied at the input
of the MAR-4 IF buffer amplifier.
The prototype's threshold of data recovery was approximately -45 dBm
at the FSK demodulator subassembly input. In general, any S-meter indication
below 20 or so corresponds to a signal level which is too low for reliable
operation. The signal path depicted in the photo
yields S-meter readings around 35-40 with the optional 100 mW Gunn diodes
installed in the M/A-Com 10 GHz transceiver modules, suggesting a usable
S/N margin of about 10 dB. (The IF strip's gain of 40 dB can be added to
the levels shown above to determine the signal available at the Gunnplexer
mixer diode for a given S-meter reading.)
Like the tuning/fault meter circuit, the exact design
of the signal-strength monitoring circuit can be left to the builder's
discretion. Omitting the S-meter circuit isn't recommended, as it is almost
indispensible for antenna aiming and link diagnostics.
Signal Level
Displayed S-meter indication
Actual RSSI voltage in mV
IF power in dBm
(No signal)
-5.0
2.4
-60
-4.6
2.8
-55
-3.6
3.8
-50
0.0
6.8
-45
8.0
15
-40
20
25
-35
28
33
-30
38
42
-25
50
53
-20
60
63
-15
65
68
-10
68
70
-5
70
72
0
72
74
Apart from the output voltages available from the main power supply,
the transceiver subassemblies require power busses at +9, +/-5, and -8
VDC. These voltages are easy to obtain from 3-terminal 78/79xx regulators
driven from the main power supply. (The +5V supply to the MC13155 is derived
from a dedicated 7805 regulator inside the FSK
demodulator subassembly). To keep the diagrams simple, none of the
3-terminal regulators and their associated bypass capacitors are shown
on the schematics, although they can be seen in the overall
view of the prototype transceiver assembly if you look closely. Two
regulators are mounted on the cover of the IF strip subassembly, with the
remaining pair mounted to either side of the 9-pin D-sub connector where
the power supply and signal-monitoring cable from the link control unit
plugs in.
Because the microwave link is a broadband system with very high gain,
good RF shielding and decoupling practices are quite important when assembling
the transceiver units. While it's not necessary to follow the prototype
examples faithfully in every respect, a few general suggestions are worth
pointing out.
Perhaps the most important advice of all is to study the manufacturer's
data sheets and app notes for the parts you plan to use, even if you don't
understand everything being discussed. The manufacturer's application
engineers have seen every problem under the sun, and it's in their interest
to warn you about potential complications that are likely to arise when
working with their chips.
This practice not only helps keep internal RF signals where they belong,
but also makes it easy to assemble and test individual modules independently
of their neighbors. Not only that, you'll have the option of reusing entire
modules in future projects, instead of salvaging individual parts! Although
it does add to the effort and expense of construction, I am a die-hard
believer in enclosing all but the most trivial RF circuits in Hammond
1590-series boxes. These enclosures are available in a variety of sizes
from almost every mail-order vendor including Digi-Key
and Mouser. The IF strip and FSK demodulator
subassemblies were built in Hammond 1590A boxes which were themselves enclosed
in a larger 1590D chassis. The FSK modulator circuit and RJ-45 jack used
for the AUI cable connection were installed in a separate 1590B enclosure
which was bolted to the exterior of the main chassis box.
Each transceiver's power and tuning/S-meter lines are brought into
the outer enclosure via a 9-pin D-sub male jack. Pairs of good RF-quality
0.1 uF ceramic capacitors and 4.7 uF tantalum capacitors are used to bypass
the +/- 12V lines at the connector itself, while the S-meter and tuning-meter
signal lines are bypassed with 0.1 uF capacitors alone. The reason for
bypassing at the point of entry, as opposed to any other point within the
enclosure, is that the length of wire between the D-sub connector and the
bypass point will act as an antenna, radiating stray RF signals from paging,
FM, and VHF television stations into the supposedly RF-tight enclosure!
Even a few centimeters of unbypassed connecting wire can couple enough
energy into the enclosure to degrade the link's sensitivity by several
dB. If either of your transceivers shows more than 5-10 millivolts at the
S meter with all enclosure covers in place and the other unit powered off,
you should suspect inadequate shielding or bypassing. An excellent tutorial
on RF shielding practices can be found at Wes
Hayward, W7ZOI's site.
Components such as the Mini-Circuits MMIC amplifiers, OPA603 opamps,
and MC13155 FSK demodulator are nothing short of miracles of solid-state
integration. A few short years ago, parts like these simply could not be
manufactured, much less purchased for the price of a nutritious lunch for
two at McDonald's. Although these three examples are devices with widely-differing
purposes and specifications, they all share a common trait which may not
be obvious at first glance: very high gain over a bandwidth extending from
DC into the hundreds of megahertz. Don't even think about mounting these
parts on conventional "perfboard" or Vector-style boards that lack an extensive
ground plane. Keep lead lengths short, and pay attention to places (e.g.,
the OPA603 input pins) where it's actually best to leave some clearance
to the ground plane in order to minimize stray capacitance. As the photographs
will reveal, my personal sympathies lie with the "dead-bug" or "ugly"-construction
school of thought, in which components are suspended by their own leads
over a ground plane at the surface of an unetched copper-clad board. Conventional
PC boards are fine as well, as long as their layouts and ground planes
are carefully designed.
In soldering to the Gunnplexer diode terminals, it's best to use a very
hot iron with a small, well-tinned tip to keep from "soaking" the diodes
with excessive heat. A low-wattage iron brings with it a greater risk of
damage to solid-state devices, since it will take longer to make a good
connection. A grounded iron suitable for use with static-sensitive devices
is ideal, if one is available.
Next, apply +5V DC to each Gunnplexer's varactor diode to establish
the voltage corresponding to the center IF frequency. It's best to feed
this voltage through a temporary network consisting of a 10K resistor followed
by a 0.1 uF bypass capacitor, to keep power-supply noise from complicating
the alignment process. If you have access to a microwave spectrum analyzer
or frequency counter, simply adjust the Gunnplexers' mechanical tuning
screws to set one unit to 10.323 GHz and the other to 10.178 GHz. Again,
the exact frequencies are not critical as long as they are within a few
MHz of 145 MHz apart. Keep track of which unit is which -- you'll need
to install the 10.178 GHz Gunnplexer into the "low-frequency" transceiver
to maintain the correct Ethernet signal polarity.
If you don't have a way to measure the Gunnplexer output frequencies
directly, you'll need to monitor the mixer output of one unit to measure
the difference in their frequencies. With both mixer-diode shorting straps
in place, carefully solder a 2.2K resistor between the mixer terminal of
one Gunnplexer and its ground terminal. The shorting strap on this unit
may now be removed. Separate the two Gunnplexers by a few inches, with
their apertures facing each other, and connect a VHF frequency counter,
spectrum analyzer, or receiver to the mixer terminal. Slowly tighten either
Gunnplexer's tuning screw into its resonant cavity until a difference frequency
of 145 MHz is received. This will appear as a quieting signal on a 2M receiver.
To make sure you're in the 10.0 - 10.5 GHz amateur band, you may wish to
tighten the screw on the other Gunnplexer at this point until the
same result is achieved. Clockwise rotation of the tuning screw will lower
the transmitted frequency. Since the Gunnplexers' initial frequency as
delivered from the factory is close to 10.525 GHz, this two-step process
should put your units safely within the ham band. Should you find your
units hopelessly misaligned, an inexpensive police radar detector with
X-band capability can help you retune them to a frequency near 10.525 GHz.
Radar detectors typically respond to a range of frequencies around 60 MHz
either side of 10.525 GHz, so you should be able to achieve a good center-frequency
response with the tuning screws.
As an alternative means of alignment, if you have access to a 145 MHz
signal source (either a 2M rig or a calibrated RF signal generator) you
may find it more convenient to build and align the FSK demodulator assemblies
first. Temporarily connect one of the demodulator assemblies to the mixer
diode test point. (The IF strip will not be needed for close-range testing.)
The output voltage between pins 4 and 5 of the MC13155 will pass through
0V as the difference frequency between the two Gunnplexers passes through
145 MHz. In fact, this makes a good sanity check even if you have already
aligned the Gunnplexers by other means. Watch out for false nulls -- the
response at the correct adjustment point should be sharp and symmetrical
to either side.
An accurate 145 MHz signal source should be used for final alignment
of each FSK demodulator circuit. This can be a 2M transmitter operating
on the bench a few inches away from the circuit under test, a calibrated
signal generator, or even an aligned pair of Gunnplexers. The latter option
can be especially attractive since a good frequency match between the Gunnplexers'
IF difference frequency and each FSK discriminator's tank circuit is exactly
what we're trying to achieve.
With the signal source turned on, use a digital VOM or oscilloscope
to monitor the differential voltage between pins 4 and 5 of the MC13155.
It's easy to connect a monitoring instrument if you route these signals
to a pair of RCA jacks, as in the prototype.
Spread or compress the coil's turns by hand, using a toothpick or other
nonmetallic alignment tool, in order to minimize the voltage difference
between the two pins. Your goal should be to come within +/- 10 to 20 millivolts
of 0V. Fine adjustment can be accomplished by physically bending the leads
of the 180-ohm resistor or 22-pF capacitor to move the component's body
back and forth relative to the coil.
Finally, remove the alignment tool and install the FSK demodulator subassembly's
cover to ensure that the coil's tuning characteristics don't change drastically
with the cover in place. If a marked change in alignment occurs, remove
the cover and separate the coil's turns slightly to compensate.
If you have a 50-ohm signal generator with a calibrated output attenuator,
you may wish to check the amplitude response at each demodulator assembly's
RSSI output against the table above. RSSI output
values within +/- 20% should be noted at each input level setting.
Links which are constructed with substantially different microwave transceiver
modules, or which need to be carefully optimized for maximum S/N performance,
should employ a variable resistor at R1 to simplify calibration. If you
have an oscilloscope with a 50-100 MHz bandwidth, it would definitely be
a good idea to verify that your LO drive waveform
is similar to the prototype's in amplitude and DC offset with actual Ethernet
data being transmitted. Since I do not have a copy of the official Ethernet
specification to work with, I'm assuming that wide variations in
AUI output amplitude are not likely to show up among different manufacturers'
hubs and network interface cards! As always, I'd appreciate any feedback
regarding this issue.
Most of these sources have already been mentioned above, but are listed here for ease of reference.
Now that we're past all the mandatory disclaimers, the reality is that as an operator of an X- or K-band data link, you are far more likely to be the victim than the cause of any interference to other radio services. Especially in urban areas, these bands are heavily populated by users ranging from supermarket door-openers, to microwave burglar alarms, to police traffic radar units -- all of which operate at frequencies and power levels which are similar to those used by the link. It is hardly an uncommon occurrence to find these users operating with unstabilized Gunnplexers which have long since drifted out of their assigned frequency slots and into the 10-10.5 GHz ham band. With good directional antennas on your link, operating near the minimum power necessary for reliable communications, the odds that you will cause harmful interference to any other service are essentially nil. However, you may need to be prepared to adjust your Gunnplexer tuning screws as necessary to move your link as far away as possible from any interfering sources. As noted above, if you can solve this type of problem by throwing more antenna gain at it, by all means do so! Like other FM radios, the microwave link will exhibit a "capture effect" which will tend to block interference from other signals arriving at the receiver with substantially less power relative to the desired carrier. A good antenna system can make all the difference when faced with interference from off-axis sources.
With the recent proliferation of personal cellular telephones and other high-frequency consumer devices, RF safety has become a topic of frequent discussion and debate. As with most such unresolved questions, good technical judgment demands that we choose a position somewhere between the Chicken Little crowd ("if it saves just one child, we've all got to freeze to death in the dark!"), and reckless disregard for known biological consequences of exposure to excess RF radiation ("hell, back on the DEW Line in Iceland we used to stand in front of 20-megawatt magnetron feeds to keep warm!"). It's well known that RF power levels high enough to cause thermal (tissue-heating) effects in humans carry a high degree of risk to organs such as the eyes, genitals, and brain. Certainly no sane person would defeat a microwave oven safety interlock and stick his/her head inside to see if the oven's working. However, several recent studies have also suggested a link between long-term exposure to low-level (athermal) RF radiation and certain forms of cancer. These newer studies are far from conclusive, having generally taken the form of epidemiological surveys which are limited to pointing out correlations, rather than causal links, between RF exposure and health risks to humans. Still, given what we know (and perhaps more important, don't know) about the biological effects of RF radiation at varying frequencies and power levels, it's not unwise to minimize long-term exposure to even the low levels of power emitted by the microwave data link's Gunnplexers.
At first glance, the power levels produced by a Gunnplexer transmitter may seem too low for even the most paranoid user to be concerned about. The power radiated from a Gunnplexer with a horn antenna is about the same as that emitted by a cellular telephone -- less than a thousandth of the RF power present in a household microwave oven. However, cell phones use vertical antennas that exhibit little or no gain over an isotropic (directionless) radiator, while microwave-link applications commonly employ horn or dish antennas with 15 to 30 dBi of gain. Such antennas can deliver a surprising amount of power to a human-sized target. By way of analogy, a 1-milliwatt laser appears much brighter than a 100-watt light bulb because the laser's "antenna" exhibits about 50 dB of gain over the relatively isotropic light bulb! Although it's easy to reassure yourself that these combinations of Gunn diode power and antenna gain are still relatively safe, using resources such as the N5XU RF safety calculator hosted by the University of Texas, it simply does not make sense to install a Gunnplexer in an application where it will be pointed directly at someone's head from only a few feet away. Treat any microwave antenna with the same respect you would pay a potentially-loaded firearm. Never peer into an operating feed horn or antenna aperture, no matter how low the power -- and don't install microwave antennas where uninformed parties (children, pets, inquisitive visitors...) are likely to do the same.
Kirkland, Washington
August 9, 1999
John, may I suggest a small change in ur schematic? Now, you are allowing the 2.2K load on the mixer to be in parallel with the 4:1 transformer. Not bad, but protection can be equal and a bit more signal found if you do it the way I indicate on the attached .pdf drawing, called mixer2.pdf...
Thanks to Alan for this suggestion. There's certainly no reason why the mixer diode's DC load resistor can't be placed in series with the matching-transformer primary.
I have corrected the error in the FSK modulator schematic, and will fix the corresponding Orcad file as soon as possible.
I had guessed what you were thinking when you put the AFC together pretty well and I think it will work, but not as well as the tweaked circuit. I'm sure neither are really optimum in any sense of the word though. The reason I still like mine a tad better is that I measured the actual varactor voltage as I heated the gunnplexor on one end up with the heat gun. For your circuit I could notice almost no change in the voltage at the collector of the 2N2222 or at the varactor diode either one. With the 2N2222 in the active region I can see it change by several volts in the right direction. It is probably the case that the 2N2222 here happened to have a higher beta than yours did, and so works not so well in your circuit. I like to think of my circuit with the 2N2222 as a voltage to current converter, and think of the current as driving the op amp. At least for me, having the BJT in the active region with its more linear behaviour makes the feedback range much greater than having it most of the time in the saturated mode. Any feedback system should work better with a higher loop gain, as long as it is still stable.
I haven't encountered any trouble of this nature with the prototype, but if you're having problems achieving robust operation of the AFC circuit in your link, give Rob's modification a try!