| United States Patent |
4,958,638
|
|
Sharpe
, et al.
|
September 25, 1990
|
Non-contact vital signs monitor
Abstract
An apparatus for measuring simultaneous physiological parameters such as
heart rate and respiration without physically connecting electrodes or
other sensors to the body. A beam of frequency modulated continuous wave
radio frequency energy is directed towards the body of a subject. The
reflected signal contains phase information representing the movement of
the surface of the body, from which respiration and heartbeat information
can be obtained. The reflected phase modulated energy is received and
demodulated by the apparatus using synchronous quadrature detection. The
quadrature signals so obtained are then signal processed to obtain the
heartbeat and respiratory information of interest.
| Inventors:
|
Sharpe; Steven M. (Atlanta, GA);
Seals; Joseph (Stone Mountain, GA);
MacDonald; Anita H. (Tucker, GA);
Crowgey; Scott R. (Avondale Estates, GA)
|
| Assignee:
|
Georgia Tech Research Corporation (Atlanta, GA)
|
| Appl. No.:
|
213783 |
| Filed:
|
June 30, 1988 |
| Current U.S. Class: |
600/407; 600/430; 600/484; 600/534 |
| Intern'l Class: |
A61B 005/02; A61B 005/08 |
| Field of Search: |
128/653,716,721,670,671,782
|
References Cited
U.S. Patent Documents
| 3483860 | Dec., 1969 | Namerow | 128/653.
|
| 3951134 | Apr., 1976 | Malech | 128/653.
|
| 4085740 | Apr., 1978 | Allen, Jr. | 128/653.
|
| 4488559 | Dec., 1984 | Iskander | 128/653.
|
| 4638808 | Jan., 1987 | Mawhinney | 128/653.
|
Primary Examiner: Howell; Kyle L.
Assistant Examiner: Pfaffle; K. M.
Attorney, Agent or Firm: Hurt, Richardson, Garner, Todd & Cadenhead
Goverment Interests
GOVERNMENT INTEREST
This invention was made with Government support under contract No. N
00014-82C-0930 awarded by the Department of the Navy and under Contract
No. F 33615-83D-0601 awarded by the Department of the Air Force. The
Government has certain rights in the invention.
Claims
What is claimed is:
1. An apparatus for measuring simultaneously the heart and respiration
rates of a subject comprising:
transmitting means for directing a beam of frequency modulated, continuous
wave radio frequency energy towards a body portion of said subject;
receiving means for receiving said frequency modulated beam as a
motion-related, phase modulated reflected signal from said body portion;
and
signal processing means for extracting the heart and respiration rates from
said phase modulated reflected signal.
2. The apparatus of claim 1 wherein the transmitting means comprises:
voltage controlled oscillator means to produce a continuous wave radio
frequency signal output;
modulator means for providing a time-varying ramp waveform to frequency
modulate said continuous wave radio frequency signal output of said
voltage controllable oscillator means;
directional coupler means to split the oscillator's output into a first
signal for transmission and a second signal for mixing with said phase
modulated reflected signal;
attenuator means to control the radiated power of said first signal output
and said second signal output; and antenna means for transmitting said
first signal output toward said body portion.
3. The apparatus of claim 2 wherein said modulator means comprises a
digital timing circuit means for providing a first reference signal and a
second reference signal, said reference signals having a relative phase
difference of 90.degree..
4. The apparatus of claim 3 wherein said modulator means further comprises
an analog ramp generator circuit means for forming the time-varying ramp
waveform used to frequency modulate said radio frequency signal output of
said voltage controllable oscillator means.
5. The apparatus of claim 3 wherein the receiving means comprises:
antenna means for receiving said phase-modulated reflected signal from said
subject;
mixing means for combining said received phase-modulated reflected signal
with said second signal output from said voltage controllable oscillator
means to produce a different signal output containing harmonic frequency
components of the time-varying ramp weigh form;
preamplifier means for amplifying and filtering noise from the different
signal output from said mixing means to produce a detected signal; and
demodulator means for synchronously detecting in-phase and quadrature
components from said detected signal.
6. The apparatus of claim 5 wherein the demodulator means comprises:
a first bandpass filter means to remove said harmonic frequency components
from said detected signal output from said preamplifier means and said
first bandpass filter means providing a first detected output signal and a
second detected output signal;
a first synchronous detector having inputs comprising said first reference
signal from said digital timing circuit means and said first detected
output signal, and the output of said first synchronous detector
comprising the in-phase component of said detected signal;
a second synchronous detector having inputs comprising said second
reference signal from said digital timing circuit means and said second
detected output signal, and the output of said second synchronous detector
comprising a quadrature component of said detected signal; and
a plurality of second bandpass filters to filter the output of each of said
synchronous detectors.
7. The apparatus of claim 6 wherein the demodulator means further comprises
a weighting circuit to suppress range sidelobes in said detected signal by
multiplying said detected signal by a weighting function to form a
weighted detected signal.
8. The apparatus of claim 7 wherein the weighting function consists of a
sinusoidal term and a DC offset term which describes the Hanning window.
9. The apparatus of claim 7 wherein the weighting function consists of a
sinusoidal term and a DC offset term which describes the Hamming window.
10. The apparatus of claim 9 wherein the weighting circuit comprises an
operational amplifier-based bi-quad lowpass filter having a first
potentiometer connected to the input of said bi-quad low pass filter to
vary the amplitude of the sinusoidal term in the weighting function, a
second potentiometer connected to the input of said bi-quad low pass
filter to generate a DC offset term, a summing junction connected to the
input of said bi-quad low pass filter to add the sinusoidal term and the
DC offset term, and a third potentiometer connected between the input and
the output of said bi-quad low pass filter to set precisely the cutoff
frequency of said bi-quad lowpass filter.
11. The apparatus of claim 6 wherein the signal processing means comprises:
a plurality of sampling means for measuring the filtered outputs of each of
said synchronous detectors at a specified rate and producing a plurality
of digital sampled outputs;
a plurality of digital filtering means for passing each of said plurality
of digital sampled outputs above a cutoff frequency to produce a plurality
of filtered digital sampled outputs; and
algorithm means for computing the autocorrelation function of the plurality
of said filtered digital sampled outputs.
12. The apparatus of claim 5 further comprising:
a single antenna means for transmitting a beam of frequency modulated,
continuous wave radio frequency energy and for receiving a phase modulated
reflected signal;
a circulator means for recovering said received phase modulated reflected
signal from said antenna means; and
isolator means to prevent leakage of said second signal output from said
voltage controlled oscillator means from reaching said antenna means.
13. A method for measuring simultaneously the heart and respiration rates
of a subject comprising the steps of:
transmitting a beam of frequency modulated, continuous wave radio frequency
energy towards a body portion of said subject;
receiving said frequency modulated beam as a motion-related, phase
modulated reflected signal from said body portion of said subject; and
processing the phase modulated reflected signal to extract the heart and
respiration rates from said phase modulated reflected signal.
14. The method claim 13 wherein the step of transmitting a beam of
frequency modulated, continuous wave radio frequency energy comprises:
producing a continuous wave radio frequency signal;
modulating said continuous wave radio frequency signal with a time-varying
ramp waveform to produce a frequency modulated, continuous wave radio
frequency signal;
splitting said frequency modulated, continuous wave radio frequency signal
into a first signal for transmission and a second signal for mixing with
said phase modulated reflected signal;
controlling the radiated power of said first signal; and
transmitting said first signal towards the body portion of said subject.
15. The method of claim 14 wherein the step of modulating further comprises
generating a pair of reference signals having a relative phase difference
of 90.degree..
16. The method of claim 15 wherein the step of receiving a motion-related,
phase modulated reflected signal comprises:
receiving said frequency modulated beam as phase-modulated reflected signal
from said subject;
mixing said phase-modulated reflected signal with said second signal to
produce a difference signal output containing harmonics of said
time-varying ramp waveform;
amplifying and filtering said difference signal output to produce a
detected signal;
demodulating said detected signal to produce in-phase and quadrature
components of said phase modulated reflected signal.
17. The method of claim 16 wherein the step of demodulating said detected
signal comprises:,
filtering said detected signal to remove from said detected signal the
harmonic frequency components;
inverting said filtered detected signal to produce a first filtered signal
and a second filtered signal, said first filtered signal and said second
filtered signal being equal in amplitude but opposite in phase;
mixing said first filtered signal with the first of said pair of said
reference signals to form an in-phase output signal;
mixing said second filtered signal with the second of said pair of
reference signals to form a quadrature output signal,
filtering the- in-phase output signal to remove DC and high frequency
mixing components to form a filtered in-phase signal; and
filtering the quadrature output signal to remove DC and high frequency
mixing components to form a filtered quadrature signal.
18. The method of claim 16 wherein the step of demodulating said detected
signal includes multiplying said detected signal output by a weighting
function to suppress range sidelobes contained in said detected signal.
19. The method of claim 18 wherein the step of processing the
phase-modulated reflected signal to extract the heart rate and respiration
rates comprises:
sampling the filtered in-phase signal at a specified rate to produce a
digital sampled in-phase output;
sampling the filtered quadrature signal at a specified rate to produce a
digital sampled quadrature output;
high-pass filtering the digital sampled in-phase output to remove low
frequency components;
high-pass filtering the digital sampled quadrature output to remove low
frequency components; and
performing an autocorrelation on the high-pass filtered digital sampled
in-phase output and the high-pass filtered digital sampled quadrature
output.
20. An apparatus for measuring diagnostic information regarding the
mechanical function of the heart, lungs, chest wall, or any anatomical
part of the body of a subject undergoing motion comprising:
transmitting means for directing a beam of frequency modulated, continuous
wave radio frequency energy towards a selected body portion of said
subject;
receiving said frequency modulating beam as means for receiving a
motion-related, phase modulated reflected signal from said selected body
portion; and
signal processing means for extracting the diagnostic information of
interest from said motion-related phase modulated reflected signal.
21. A method for measuring diagnostic information regarding the mechanical
function of the heart, lungs, chest wall, or any anatomical part of the
body of a subject undergoing motion comprising:
transmitting a beam of frequency modulated, continuous wave radio frequency
energy towards a selected body portion of said subject;
receiving said frequency modulated beam as a motion-related, phase
modulated reflected signal from said selected body portion of said
subject; and
processing the phase modulated reflected signal to extract the diagnostic
information of interest from said phase modulated reflected signal.
Description
BACKGROUND OF THE INVENTION
This invention relates in general to the use of radar techniques to detect
minute body movements which are associated with cardiac and respiratory
activity. The invention is based on the principle that breathing and
heartbeat produce measurable phase changes in electromagnetic waves as
they reflect off of a living person. The invention offers significant
advantages over other similar and earlier approaches, including greater
sensitivity, lower radiated power, improved reliability and lower cost.
Functionally, the non-invasive, electromagnetically-based Vital Signs
Monitor (VSM) is an extremely sensitive motion detection system capable of
detecting small body motions produced by respiratory and cardiac
functioning. Motion detection is achieved by transmitting an interrogating
electromagnetic field at the target of interest, and then measuring the
time-delay of the return signal reflected back from the surface of the
target. When the target surface is moving, as does the surface of the
chest in conjunction with respiratory and cardiac activities,
corresponding variations will be observed in the measured time delay. The
observed variations can be used to determine motion-related target
parameters such as displacement and velocity.
In the medical field, it is essential that a subject's respiration and
heartbeat be capable of being measured. The medical profession is
accustomed to voltage-derived electrocardiogram waveforms for monitoring
heartbeat. Most respiration monitors also require physical connection to
the subject's body. Many commercially-available devices are available for
measuring heart and respiration rates, but most of them are
electrode-based requiring physical contact with the subject. Devices
requiring physical contact, however, are difficult to use on children
susceptible to sudden infant death syndrome (SIDS) or burn patients who
cannot tolerate the touch of electrodes. Many infants wear sensors while
they sleep that trigger an alarm if their breathing stops, but electrodes
attached to the child can be jarred loose as the infant tosses and turns.
The invention has similarities with motion-detection systems based on
ultrasonic or optical techniques. However, an electromagnetically-based
approach offers several advantages for monitoring of vital signs-related
motions. For example, with proper antenna design, an interrogating
electromagnetic field will suffer minimal attenuation while propagating in
air (unlike ultrasonic signals which propagate poorly in air). Thus, the
electromagnetically-based Vital Signs Monitor can easily be used in a
completely non-contacting mode and can, in fact, be placed an appreciable
distance from the test subject if required. Electromagnetic signals in the
microwave band are also capable of penetrating through heavy clothing.
This offers advantages over optical techniques which would have a
difficult time of detecting motion through even thin clothing. Another
feature of an electromagnetically-based approach is that the system could
be designed to simultaneously interrogate the entire chest surface and
provide information pertaining to any respiratory or cardiac function
manifested as chest wall motions. Conversely, by modifying the antenna
design, a localized region of the chest surface could be interrogated to
obtain information about some specific aspect of respiratory or cardiac
function. Such versatility would be difficult to achieve with other motion
detection techniques.
In the prior art the patent to Allen, U.S. Pat. No. 4,085,740 discloses a
method for measuring physiological parameters such as pulse rate and
respiration without electrodes or other sensors being connected to the
body. A beam of electromagnetic energy is directed at the region of
interest which undergoes physical displacement representing variations in
the parameter to be measured. The phase of the reflected energy when
compared with the transmitted energy indicates the amount of actual
physical movement of the body region concerned. The method does disclose
simultaneous detection and processing of respiration and heart beat;
however, frequency modulation is not used, therefore and the subject must
be reasonably still. The receiver includes two channels and in one of them
the received signal is mixed with a signal substantially in quadrature
with the transmitted signal to maximize amplitude output in those cases in
which the received signal is 180.degree. out of phase with the transmitted
signal.
The patent to Kaplan, et al., U.S. Pat. No. 3,993,995 discloses an
apparatus for monitoring the respiration of a patient without making
physical contact. A portion of the patient's body is illuminated by a
transmitted probe signal with the reflected echo signal detected by a
monitor. The phase difference between the transmitted and reflected
signals is determined in a quadrature mixer which generates outputs
indicative of the sine and cosine of the difference signal. These two
outputs are coupled to differentiators and when both time derivatives are
substantially zero an x-ray unit is triggered since it represents an
instant of respiration extrema (apnea). The outputs of the quadrature
mixer are also coupled to a direction of motion detector which indicates
inhalation or exhalation.
The patent to Kearns, U.S. Pat. No. 4,289,142 discloses a respiration
monitor and x-ray triggering apparatus in which a carrier signal is
injected into the patient's thorax which is indicative of the
transthoracic impedance of the patient. This impedance changes as a
function of the respiration cycle. The carrier signal is injected through
electrodes coupled to the patient's thorax. The transthoracic impedance
has an alternating current component having a respiratory component
between 0.2 to 5 ohms and a cardiac component varying between 0.02 to 0.2
ohms.
The patent to Robertson et al., U.S. Pat. No. 3,524,058 discloses a
respiration monitor which uses body electrodes to direct an electric
current to a particular part of the patient's body where changes in
electrical impedance provide output signals that vary with respiration.
The patent to Bloice, U.S. Pat. No. 3,796,208 discloses an apparatus for
monitoring movements of a patient including a microwave scanner (doppler
radar) which creates a movement sensitive field surrounding part of the
patient. Movements of the patient create disturbances in the field which
are monitored and which trigger alarm circuitry.
Also in the prior art, apexcardiograms (ACG), which represent a contact
technique for measuring small chest surface motions overlying the cardiac
apex, have been used to estimate cardiac contractility, left ventricular
end-diastolic pressure, pressure changes during atrial systole and cardiac
ejection fraction, in addition to diagnosing myocardial wall abnormalities
and dysfunction. One of the problems associated with the use of an ACG for
the estimation of cardiac function is that the motions recorded are
indicative only of activity at the apex of the heart and not of the heart
as a whole. Analysis of the VSM waveform is potentially a better choice
for estimation of cardiac function since the larger beamwidth of the VSM
antenna actually integrates motion over a certain area of the chest. In
addition, since the VSM waveform appears to contain information related to
aortic and other vascular pulses, it can be used to measure pulse transit
times directly out of the heart into the aorta. This measurement can
potentially be used as a non-invasive, non-contact means of estimating
blood pressure as discussed by L. A. Geddes, M. Voelz, C. F. Babbs, J. D.
Bourland and W. A. Tucker in "Pulse Transit Time as Indicator of Arterial
Blood Pressure," Psychophysiology, Vol. 18, No. 1, pp. 71-74, 1981. This
paper showed that the pulse-wave velocity in the dog aorta increased
linearly with increasing diastolic pressure. Similarly, pulse pressures
may be related to either the magnitude of the aortic peak in the VSM
waveform, or possibly to the rate of rise of this peak.
SUMMARY OF THE PRESENT INVENTION
It is therefore an object of the present invention to Provide an
electromagnetic vital signs monitor that can reliably measure
simultaneously both heart and respiration rates.
It is a further object of this invention to provide a device for measuring
physiological parameters without physically contacting the subject with
sensors or like attachments.
It is a further object of this invention to provide a device for measuring
physiological parameters of subjects remotely at distances up to
approximately 20 feet.
It is a further object of this invention to provide a device for
non-contact and non-invasive diagnosis and monitoring capabilities of
cardiac, pulmonary, and thoracic mechanical functions resulting from
normal or induced physiological responses, trauma, disease or response to
therapy.
It is a further object of this invention to provide a device for measuring
remotely the physiological parameters of subjects that are fully clothed
and that can be either stationary or moving while sitting or standing.
It is a still further object of this invention to provide a device which
can be used as an apnea monitor for patients in hospital or clinic
intensive care units, or as a patient monitor in burn or trauma clinics or
in nursing homes.
It is a still further object of this invention to provide a portable device
that can be taken into patient areas for the purpose of measuring heart
beat and respiration rates.
It is a still further object of this invention to detect the presence of
persons in visually obstructed areas or under debris resulting from
certain disasters.
The non-contact electromagnetic vital signs monitor is comprised of a
coherent, linear, frequency modulated continuous wave radar with
refinements to optimize the detection of small body movements. The
transmitter of the device is frequency modulated by a linear ramp derived
from a master clock. The transmitted signal is fed to the radio frequency
(RF) network which routes a portion of the energy to the antenna which
then interrogates the subject. Signals reflected by the subject containing
motion-related phase modulation are intercepted by the antenna and applied
to the RF network where they are mixed with a portion of the original
signal.
The mixing process produces a difference signal which contains harmonics of
the original modulating ramp. Each harmonic line is surrounded by
sidebands which are related to the body movements. The relative levels of
these sidebands are a function of target range, transmitter frequency
deviation, and harmonic number. By properly choosing these latter two
parameters, signals from a desired range can be detected while others are
suppressed. The process is further refined to result in more ideal range
discrimination by multiplication by a weighting function synchronized to
the ramp which reduces the range sidelobes.
The final synchronous demodulation is accomplished by mixing the received
signal (after weighting) with both the in-phase and quadrature components
of the desired harmonic of the modulating ramp which is generated by a
synthesizer. After recovery of the in-phase and quadrature components of
the received signal, sophisticated digital signal processing can be
economically applied since the bandwidths are relatively low. In the
preferred embodiment, a high order linear phase finite impulse response
digital filter is used on each channel to reduce the dominance of the
strong respiratory signal. A complex autocorrelation is performed from
which the rates of interest may be calculated.
Still other objects, features and attendant advantages of the present
invention will become apparent to those skilled in the art from a reading
of the following detailed description of the preferred embodiment, taken
in conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of the VSM.
FIG. 2 is a functional diagram of the modulator-receiver/demodulator
without a weighting circuit.
FIG. 3 is a functional diagram of the modulator-receiver/demodulator with a
weighting circuit.
FIG. 4 is a schematic diagram of the weighting circuit.
FIG. 5 is a schematic diagram of the chopper-based synchronous detectors
used in a demodulator.
FIG. 6A is a power spectrum for a composite VSM output displaying
respiratory and cardiac spectral components.
FIG. 6B is a power spectrum for a composite VSM output using logarithmic
amplitude and linear frequency scale to demonstrate harmonic structure of
cardiac information.
FIG. 7A is a VSM return signal from a stationary subject immediately after
sampling.
FIG. 7B is a result of applying a Fast Fourier Transform to the sampled
data.
FIG. 7C is a result of applying a digital high pass filter to the VSM
return signal.
FIG. 7D is a complex autocorrelation of the filter signals.
DETAILED DESCRIPTION OF THE INVENTION
The apparatus which is used to carry out the method of the present
invention is referred to hereinafter as the VITAL SIGNS MONITOR (VSM).
At the electromagnetic frequencies of 3 and 10 GHz that have been used in
the VSM, the surface of the body is highly reflective to incident
electromagnetic fields. In addition, biological tissue is very lossy at
these frequencies and there is minimal penetration of radiated
electromagnetic energy into the body. Therefore, a return signal from a
radiated electromagnetic field incident on the body will primarily contain
information associated with events occurring at the body surface.
Motion of a target with an electromagnetically-reflective surface can be
detected by transmitting an interrogating signal at the target surface,
and then measuring the motion related time-delay of the return signal that
reflects back from the target. The interrogating signal travels at the
speed of light and the time delay experienced by the return signal is
equal to the round-trip distance to the target surface, divided by the
speed of light. Thus, the time delay of the return signal is proportional
to the range or distance to the target surface. If the target is moving in
a manner that varies the target range, variations in the measured time
delay can be used as a measure of target motion.
A simple mathematical model can be used to describe this motion detection
phenomenon. Assume that a sinusoidal interrogating signal, vi(t), of the
following form is transmitted at the target of interest,
vi(t)=A sin(wt). (1)
The return signal, vr(t), reflected from the target surface can be
represented as,
vr(t)=kA sin w(t+T). (2)
The parameter k represents losses due to propagation attenuation and
imPerfect reflection from the target surface. The parameter T represents
the time delay information of interest. To extract this information, a
portion of the transmitted signal vi(t) can be used as a reference signal
to demodulate the return signal vr(t). By combining vi(t) and vr(t) and
passing them through a nonlinear device such as a mixer or square-law
detector, an output signal vo(t) of the following form can be produced,
vo(t)=1/2 k cos(wT). (3)
The parameter k' includes the previously defined parameter k as well as
conversion losses associated with devices (splitters, couplers, mixers,
etc.) involved in the demodulation process.
Assuming that k' is relatively constant for small target motions, the
motion-related information is contained within the argument (i.e., wT) of
the cosine function. R(t) is defined as the range to the target surface
(the range is time-varying because of the target motion). As previously
noted, T is then equal to 2R(t)/c, where c is the speed of light. Using
this relationship and noting that 2 pi c/w equals the wavelength (denoted
as L), Equation (3) can be expressed in the form,
vo(t)=1/2 k' cos(4 pi R(t)/L). (4)
This result provides insight into the operation and potential problems of
electromagnetically-based motion detection. One useful observation is that
only the cosine of the angular (phase) information of interest is
accessible. Thus, for accurate recovery of the desired motion information
(i.e., R(t)), both the nominal distance to the target and the magnitude of
the target motion must fulfill certain requirements. The nominal target
distance should be set to insure that the average value of the phase term
in Equation (4) is an odd integer multiple of one-half wavelength. This is
equivalent to requiring that vi(t) and vr(t) be in phase.
The motion magnitude should also be small enough to insure that the
motion-related variations in the phase term do not exceed approximately
.+-.45.degree.. This limits operation to portions of the cosine function
that approximate a straight line and insures that Equation (4) produces a
nearly linear approximation of the motion R(t). This requirement is
fulfilled if the target motion is small in comparison to the wavelength of
the interrogating electromagnetic field. From Equation (4), it can be
deduced that the magnitude of the target motion should not exceed
approximately .+-. one-eighth wavelength. The requirements posed by
Equation (4) result from the fact that detection systems based on the
preceding model are not true coherent systems capable of providing both
phase and amplitude information. This problem can be eliminated by
splitting vr(t) into two signals. One of these signals is demodulated
against vi(t) to produce the result obtained in Equation (4). The second
signal is demodulated against a signal in phase quadrature with vi(t) to
produce a result of the form 1/2k'sin(wT). From trigonometric
relationships, the sine and cosine terms can then be used to directly
determine the desired phase information.
The use of quadrature channels eliminates the previously discussed
limitations on the target motion R(t) and development of such a capability
has been a focal point of this invention.
Turning now to the figures in which like numerals denote like parts, a
specific embodiment of the basic design for the VSM is shown in FIG. 1.
Although there are no specific operating frequency limitations on the VSM,
systems operating at frequencies of 3 GHz and 10 GHz have been implemented
so far. Thus for these specific implementations each RF section of VSM 10
includes: (1) a voltage controllable microwave oscillator 12 to produce a
frequency modulated RF signal, (2) a directional coupler 16 to split the
voltage controlled microwave oscillators output, (3) fixed attenuators 14
to control the radiated power and local oscillator level 15, (4) an
antenna 20 for transmitting the interrogating field and receiving the
target return signal, (5) a circulator 18 to recover the return signal
from the antenna, (6) a double balanced mixer 24 for demodulating the RF
return signal to obtain an IF signal (the receiver/demodulator performs
another demodulation to retrieve the phase information from the IF
signal); (7) an isolator 22 to prevent local oscillator (LO) level 5 to RF
17 leakage through the double balanced mixer 24 from reaching the antenna
20 and (8) a preamplifier 26 to minimize noise problems. In addition, a
coaxial low-pass filter is placed on the mixer IF in the 3 GHz system to
block LO to IF leakage. Other configurations of hardware components can be
used to mix the transmitted and return signals to produce the IF result;
however, the embodiment described here is the best mode currently known to
the inventors.
Frequency modulating the voltage controlled microwave oscillator 12 is an
effective and convenient way to reduce the effects of low frequency
semiconductor noise. With frequency modulation, the motion-related
information of interest that is output by the double balanced mixer 24
appears as sidebands centered at the modulating frequency. By using a
modulating frequency of several kHz, the largest portion of the
low-frequency flicker noise is avoided and greater receiver sensitivity is
achieved.
The modulator 28 provides a time-varying waveform that can be used to
frequency modulate the voltage controlled microwave oscillator 12 in the
RF section of the VSM. The main advantage of frequency modulating the
voltage controlled microwave oscillator 12 is that it makes it possible to
achieve greater receiver sensitivity, which in turn, enables the use of
lower and thus safer, radiated power levels.
The modulator 28 can be conveniently divided into two sections: a digital
section 32 which is used to provide all timing information, and an analog
section 34 which is used to form the actual ramp waveform used as the
modulating signal. A functional diagram is shown in FIGS. 2 and 3. An
operational amplifier-based Howland integrator in ramp circuit 36 is used
to create the basic ramp waveform from inputs from a oneshot 55 and a
voltage divider 38. This particular integrator is used to provide a simple
means of generating a nonlinear ramp to compensate for the nonlinear
tuning curve of the voltage controlled microwave oscillator 12, thereby
producing a relatively linear frequency ramp. Alternative ramp generator
circuits could be used as well. A transistor in ramp circuit 36 controls
the charging of the integrator. When the input to the transistor is zero
volts, the transistor cuts off and a capacitor in ramp circuit 36 charges
to 4.5 V (the reference voltage which is the output 39 of the voltage
divider 38). The current that charges the capacitor consists of two parts:
a constant current produced by the reference input voltage and input
resistor, and a variable current which is the ratio of the voltage being
amplified by the operational amplifier and a resistance R which is a
function of both the input resistor and a variable resistor. The variable
resistor is adjusted so that R may be either positive or negative. The
result is that the ramp may be expressed in the form
V(t)=V.sub.0 (1-exp(-t/RC)) (5)
where V.sub.0 is a constant. When R is positive, the ramp bows upward; and
when R is negative the ramp bows downward. A high input (+5 V) to the
transistor saturates the transistor and rapidly discharged the capacitor,
resetting the ramp.
The combination of the transistor and the Howland integrator in this
particular embodiment produces a basic modulating ramp waveform. However,
additional circuitry controls the amplitude and offset level of the
modulating waveform and compensates for nonlinearities of the voltage
controlled microwave oscillator 12. The ramp amplitude controls the total
frequency range over which the voltage controlled microwave oscillator 12
is modulated. The offset level controls the center frequency. At the
output of the Howland integrator, a potentiometer acting as a voltage
divider controls the amplitude of the ramp. The offset level of the ramp
waveform is controlled by a summing network 50 that can be used to add or
subtract a DC level to the output of the integrator as can be seen in FIG.
2. The ability to control the ramp amplitude and offset level is important
to insure that the modulating frequency is centered at an appropriate
value and to insure that operation is performed over a suitable linear
portion of the voltage controlled microwave oscillator's 12 tuning
response.
The digital timing circuit 32 generates the signal required to control
charging and discharging of the integrator as can be seen in FIG. 2. In
the preferred embodiment, a 1 MHz crystal oscillator 40 is used as a
stable timing reference. The output of this crystal oscillator 40 is input
to a binary ripple counter 42 that divides the 1 MHz crystal frequency by
a factor of 2.sup.n, where n is an integer between 6 and 9. The resultant
signal from the ripple counter 42 is then fed into a two-stage walking
ring counter 44 where two significant operations occur. The walking ring
counter 44 provides an additional division of four in frequency. The
output of the walking ring counter 44 is at the modulating frequency,
which is one of the following: 4 kHz, 2 kHz, 1 kHz, or 500 Hz. In
addition, the walking ring counter 44 outputs two periodic square waves on
lines 46, 48 that are exactly in phase quadrature (relative phase
difference of 90.degree.). These quadrature signals are essential to the
synchronous detectors 58 in the receiver/demodulator 30 which provide the
VSM 10 with a quadrature channel capability.
The VSM 10 employs a receiver consisting of a front-end double-balanced
mixer 24 and a receiver/demodulator that serves as a narrow band detector
by demodulating the IF signal to obtain the phase information; this is the
second demodulation performed in the VSM. With the FM-CW approach used in
the VSM 10, motion-related information output from the mixer appears as
sidebands centered around a carrier frequency of 4 kHz. The carrier
frequency is an integral multiple of the modulating frequency. An
appropriate narrow band detection scheme is required to extract these
motion-related sidebands.
The receiver/demodulator 30 represents the required detection scheme and
consists of a low noise preamplifier 26 that provides needed gain and
filtering at the carrier frequency, an optional weighting circuit 52 to
attenuate extraneous returns, an inverting amplifier 54, a band pass
filter 56 to remove unwanted frequency components (especially third order
harmonics) prior to synchronous detection, a pair of synchronous
detectors, shown within block 58, that enable coherent detection of both
amplitude and phase information, and a pair of band pass filters 60 to
remove unwanted frequency components in the outputs of the synchronous
detectors (DC components and high frequency mixing products). A block
diagram of the receiver/demodulator 30 is contained in the lower half of
the VSM block diagrams in FIGS. 2 and 3.
Use of the weighting circuit 52 improves the ability of the ramp-based
frequency modulation technique used in the VSM 10 to reject extraneous
return signals. With this frequency modulation technique, the demodulation
outPut of the VSM 10 contains sinusoidal bursts due to returns from the
subject being evaluated. These sinusoidal bursts repeat at a rate equal to
the modulating frequency and the frequency of these bursts is a function
of the subject's distance from the VSM 10 (range). However,
discontinuities exist at end points of each sinusoidal burst due to
recycling of the modulating ramp. These discontinuities cause the
demodulated return signal to have a spectrum containing harmonics of the
modulating frequency. The undesired harmonic components, also know as
range sidelobes, enable return signals from other subjects to generate
frequency components at the receiver frequency.
Weighting functions substantially reduce the effects of these
discontinuities. To achieve range sidelobe suppression in this manner, a
demodulated return signal is multiplied by a weighting function which is
synchronized with a sinusoidal burst in the demodulated return signal.
Weighting functions are usually bell-shaped and have a value of unity at
their center with ends that taper to a small value. The resulting product
of a demodulator return signal in the selected window still contains
discontinuities, but the weighting procedure reduces their significance
and results in a more ideal spectrum.
The VSM 10 may be configured with or without weighting. As shown in FIG. 2,
when weighting is not used, the ramp frequency equals the, receiver
frequency, 4 kHz. FIG. 3 shows that, when weighting is used, the ramp
frequency equals 500 Hz, 1 kHz or 2 kHz. The weighting circuit 52 is shown
in FIG. 4. The weighting circuit 52 has been designed so that the detected
signal 27 from the preamplifier 26 may be multiplied by one of two
weighting functions, the Hamming window or the Hanning window. Other
windows could be chosen and the two that have been implemented are not a
limitation of the VSM 10. The equations for each of the windows are given
below:
Hanning:
w(t)=0.50-0.50 cos(2 pi t), 0<t<1 (6)
Hamming:
w(t)=0.54-0.46 cos(2 pi t), 0<t<1 (7)
Pulses at the modulating frequency are input into the operational
amplifier-based bi-quad low pass filter 62. Alternatively, many other low
pass filter designs can be implemented to achieve the same result. The low
pass filter 62 attenuates the higher ordered harmonics of the pulses,
generating a sinusoidal signal. The potentiometer 90 at the input varies
the input signal level so that the amplitude of the sinusoidal term in the
weighting function can be varied depending on the choice of weighting
function. A summing junction 91 allows a DC offset determined by another
potentiometer 92 to be added to the sinusoidal term and generates the
desired weighting function. The remaining potentiometer 92 involved with
the generation of the weighting function sets the cutoff frequency of the
low-pass bi-quad bandpass filter 56 precisely. The transistor acts as a
voltage-to-current converter, supplying a current proportional to the
weighting function to the CA3080 transconductance amplifier 64. The CA3080
64 essentially multiplies the weighting function by the signal output 27
from the preamplifier. The output of the weighting circuit 52 is then
input to the receiver 30.
The bi-quad bandpass filter 56 provides additional rejection of DC and low
frequency components and also suppresses any harmonics of the modulating
frequency that are generated by the double-balanced mixer 24. The bi-quad
design permits the cutoff frequency of the filter 56 to be tuned by
adjustment of a single feedback transistor and has the unique
characteristic of maintaining a constant absolute bandwidth as it is
tuned. The inverting amplifier stage used in the bi-quad bandpass filter
56 Provides two signals identical in amplitude but opposite in phase.
These signals each have a frequency of 4 kHz and contain low frequency
side bands corresponding to the motion-related information of interest.
The availability of the out-of-phase signals enable synchronous detection
using a conveniently implemented chopper approach.
A model of a chopper-based detector 58 is shown in FIG. 5. Because of its
out of phase outputs, the bi-quad bandpass filter 56 is represented as a
center tapped transformer 68. The chopper (electronic switch) 70 is
switched between the two outputs of the transformer 68 at a frequency of 4
kHz. Since the chopper frequency is identical to that of the information
from the transformer 68, the sum and difference frequency terms outputted
by the chopper 70 contain a DC term (difference frequency component) and a
8 kHz term (sum frequency component). The low pass filter 72 following the
chopper is used to remove the undesired sum frequency component. The
remaining DC term is dependent on the phase difference between the
information signal (55 or 57) into the transformer 68 and the reference
signal (46 or 48) controlling switching of the chopper.
For the demodulator 30 in the VSM 10, a synchronous detector 58 is
employed. A model of the detector 58 is synchronous to the model shown in,
FIG. 5. The reference signals 46 and 48 into the synchronous detector 58
have a phase difference of 90.degree. (the quadrature-phase reference
signals are provided by the walking ring counter 44 in the, modulator
subsystem). Thus, the one of the outputs 71 from the synchronous detector
58 can be considered an in-phase or I channel term that is equivalent to
the cosine of the phase of the information signal. The other output 73,
from the synchronous detector 58 can be considered a quadrature or Q
channel term equivalent to the, sine of the information signal. The I and
Q signals, 71 and 73, can be evaluated jointly or independently to extract
the motion-related information of interest.
The phase of the demodulated return signal varies as a function of target
motion. Therefore, the I and Q signals, 71 and 73, that are output by the
synchronous detector 58 are not true DC terms. Instead, these signals
occupy a frequency band related to that of the motion being detected.
Since the VSM 10 was designed to provide an almost linear estimate of
target motion, the frequency band of the I and Q channels is the same as
that of the respiratory and cardiac motions, being, detected. Thus,
filters 60 used on the outputs, 71 and 73 of the synchronous detector must
have sufficient bandwidth to pass information in the respiratory and
cardiac bands but must be narrow enough to reject unwanted noise and
mixing products.
Lowpass filters 72 and bandpass filters 60 with passbands of approximately
0.1-75 Hz are used to filter the outputs 71 and 73 of the synchronous
detector 58. The 0.1 Hz lower frequency cutoff of filter 60 effectively
blocks DC terms but is low enough to pass slow respiratory information.
The 75 Hz low-pass cutoff frequency filter 72 blocks the 8 kHz sum
frequency term generated by the synchronous detectors 58 but is high
enough to pass fast cardiac motion. In addition, the 75 Hz cutoff should
make it possible to determine if vibrations associated with respiratory
and cardiac sounds can be detected. Cascaded together, the filters 72, 60
have such a wide passband that for all practical purposes, the data output
may be considered unprocessed. Therefore, suitable signal processing
techniques must be used to extract useful information from the output
data.
In order to digitally process the output data, the I and Q channels are
sampled at a rate of 100 Hz by a sampler included in the Digital Sampling
and Processing block 80, shown in FIG. 1. Prior to sampling, the I and Q
channels are input to a pair of 34 Hz second order low pass anti-aliasing
filters, which are included in digital sampling block 80 in FIG. 1.
Without these anti-aliasing filters, it would be possible for signal
components over 50 Hz, half the sampling rate, to take on the identity of
lower frequencies, resulting in signal distortion.
Typically, respiratory rates of normal subjects correspond to frequencies
of approximately 0.12-0.30 Hz (7-18 breaths per minute) while cardiac
rates correspond to approximately 0.8-1.5 Hz (48-90 beats per minute).
Since there is more than an octave difference between the highest
respiratory frequency and the lowest cardiac frequency, it is possible to
examine the individual respiratory and cardiac components.
FIG. 6A shows a power spectrum for a composite VSM return signal obtained
with a laboratory signal analyzer. The linear amplitude and logarithmic
frequency scales permit detailed evaluation of the respiratory spectrum.
The fundamental respiratory component has an RMS amplitude of 280
millivolts while the fundamental cardiac component has an RMS amplitude of
only 17.9 millivolts. The respiratory component occurs at 0.21 Hz
corresponding to a respiratory rate of approximately 12 to 13 breaths per
minute. The smaller cardiac spectral component occurs at 1.18 Hz
corresponding to the subject's heart rate of approximately 70-71 beats per
minute. Harmonics of the respiratory fundamental can also be observed at
0.42 and 0.63 Hz. The second harmonic of the respiratory motion is
approximately 9 dB lower than the fundamental respiratory component. The
third harmonic is approximately 21 dB below the fundamental respiratory
component. Knowledge of the relative strengths of the individual harmonic
components can be used to identify specific features in the corresponding
time based signal. Such information can be useful for characterizing or
classifying the breathing pattern. By changing to a more convenient
logarithmic amplitude scale and a linear frequency scale, it is possible
to better evaluate the cardiac spectrum as shown in FIG. 6B. In this
figure, wider frequency resolution has resulted in a smearing of the
respiratory portion of the spectrum so that only a single respiratory
component appears at a frequency of approximately 0.15 Hz. It can be seen
that the fundamental, second, third, fifth and possibly sixth harmonics
are strong while the fourth harmonic is partially suppressed. Again,
knowledge of such spectral information is valuable for evaluating aspects
of cardiac function. One of the difficulties with spectral analysis is
that the rates determined actually represent the average rate of several
events. With average but not instantaneous rate values available, it is
difficult to compute useful parameters such as heart rate variability.
If cardiac and respiratory functions are to be evaluated from familiar
time-based waveforms, filtering techniques capable of separating the
composite signal from the VSM 10 into its individual respiratory and
cardiac components are required. In practice, it is relatively simple to
build filters to extract the desired respiratory component while several
factors combine to make recovery of cardiac information difficult to
achieve. For example, for most test subjects the peak-to-peak levels of
the respiratory-related signal is a factor of 5-20 times greater than that
of the corresponding cardiac-related signal. Thus, filters used to recover
cardiac information in the presence of a strong respiratory signal must
greatly attenuate the low-frequency band occupied by the respiratory
information. This requires filters with a sharp-rolloff (i.e., a large
number of poles). Because of the potentially small separation between the
respiratory and cardiac information bands, it is necessary that a filter
used to recover cardiac information have a precise cutoff frequency. In
addition, since respiratory and cardiac rates will vary significantly, it
is necessary that the filter's cutoff frequency be adaptive or at least
adjustable. The limited capabilities of simple analog filters appear
inadequate for this task. Thus, a digital signal processing capability is
required for the VSM to achieve reliable rate determination and to produce
diagnostically useful respiratory and cardiac waveforms. What follows is a
description of the signal processing methods used in the VSM, but those
skilled in the art will understand alternative implementations are
possible.
In order to determine the cardiac rates, the analog I and Q signals are
first sampled at a 100 Hz rate and then high pass filtered digitally with
a cutoff frequency between 0.75 and 1.0 Hz. These filters, though not
shown individually, are included with the Digital Sampling and Processing
block of FIG. 1. The complex autocorrelation to the two filtered signals
is then computed, as shown in FIG. 1 by the Autocorrelation block 82.
Periodicities in the signal due to cardiac related motions appear as
relative maxima in the real part of the autocorrelation function. The
signal processing is not performed in real time.
Digital signal processing techniques can be divided into two types: time
domain techniques and frequency domain techniques, both of which are well
known in the art of signal processing. With frequency domain techniques,
the digitized I and Q channel signals are transformed into the frequency
domain, either with a Fast Fourier Transform (FFT) or some non Fourier
spectral estimation algorithm, and a peak detection algorithm then
estimates the frequency of the spectral line which corresponds to the
cardiac rate. Time domain techniques require the processing of time domain
signals with filtering and autocorrelation algorithms to obtain other time
domain signals in which the time period between relative maxima can be
more easily detected than in the original signal. Time domain techniques
require less processing time to obtain an initial estimate of heart rate
and, therefore, are preferred to frequency domain techniques. FIGS. 7A-D
illustrate the process of heart rate termination. The I and Q channels
from the output of the VSM are shown in FIG. 7A. These two signals are
transformed into the frequency domain using a Fast Fourier Transform (FFT)
algorithm. The resultant signals are depicted in FIG. 7B. FIG. 7C shows
the I and Q channel signals after they are digitally high pass filtered.
The cutoff frequency is 1 Hz. The complex autocorrelation of the filter
signals is shown in FIG. 7D. The real part of the complex autocorrelation
is used to calculate the rates displayed at the bottom of the figure. The
digital filtering algorithm employed is the window technique. More complex
filter design procedures can achieve better filters for a given filter
length. More specifically, these more complex filter designs can yield
filters with similar performance to the filters used to obtain the data in
FIG. 7C, but the filter length could be reduced substantially, therefore
decreasing computation time.
Comparison of FIGS. 7C and 7D reveals that the autocorrelation process
yields a signal more suitable for a peak detection algorithm. The
autocorrelation algorithm used is based on a method described by G.
Hoshal, M. Siegel and R. Zapp in "A Microwave Heart Monitor and Life
Detection System," published in 1984 in IEEE Frontiers of Engineering and
Computing in Health Care on pp. 331-333. The method described by Hoshal et
al. is incorporated by reference herein. Each of the eight autocorrelation
segments shown in FIG. 7D was computed for lags between 60 and 120. Since
the sampling frequency is 100 Hz, the sampling period is 10 milliseconds,
and an autocorrelation lag of 60 corresponds to a time delay of 60 (10
ms)=0.6 seconds, or to a heart rate of 1.67 Hz or 100 beats per minute.
Similarly, an autocorrelation lag of 120 corresponds to a heart rate of 50
beats per minute.
Four hundred samples of the filtered VSM output are used to compute each of
the 8 autocorrelation segments. As a result, four seconds of data are
required to compute each heart estimate shown in FIG. 7D. If the number of
samples is too large, the aperiodic nature of the heart beat signal tends
to flatten the autocorrelation. If the number of samples is too small, the
record might not include an entire heart beat cycle. Each successive
autocorrelation calculation starts at 100 points (corresponding to one
second) further in the data record. The contrast between 7C and 7D clearly
demonstrates the ability of the autocorrelation to enhance the presence of
the heart beat signal.
While the invention has been described with respect to a preferred physical
embodiment constructed in accordance therewith, the same is by way of
example only and is not to be taken by way of limitation. It will be
apparent to those skilled in the art that various modifications and
improvements may be made without departing from the scope and spirit of
the invention. For example, while specific frequencies or frequency ranges
have been given for various components within the VSM 10 it will be
evident to those skilled in the art that other embodiments can be
developed employing different frequencies or frequency ranges. Thus,
specific operating frequencies of the voltage controlled microwave
oscillator 12, crystal oscillator 40 filters 56, 60, 72 are not
limitations of the VSM 10. Similarly, the VSM is not limited by the type
of ramp circuitry 36, weighting circuitry 52, liquid low pass filter 62,
or synchronous detection circuitry 58 disclosed herein. Other digital
sampling rates may also be used with the VSM 10. Furthermore, while the
present system has been described as applicable for the non-contact
measurement of heart rate and respiratory rate, it will be evident to
those skilled in the art that the present system may be used to measure
other types of potentially useful diagnostic information concerning
cardiac and respiratory mechanical functions which can be derived from the
VSM waveform by appropriate signal processing. Accordingly, it is to be
understood that the invention is not limited by the specific illustrative
embodiment, but only by the scope of the appended claims.
* * * * *